Regulation of electrical generator output

ABSTRACT

To regulate the electrical output of a generator, a signal is received indicative of at least one characteristic of the electrical output. A first, relatively fast-response sub-controller is configured to provide a first control signal on the basis of the at least one characteristic and a second, relatively slow-response sub-controller is configured to provide a second control signal on the basis of the at least one characteristic. An output provides a combined control signal to adjust the electrical output based on the first and second control signals.

TECHNICAL FIELD OF THE INVENTION

The invention concerns a regulator for the electrical output of agenerator, in particular wherein the generator comprises a Stirlingengine (especially a linear free piston Stirling engine with linearalternator) and a method of regulating the electrical output of such agenerator.

BACKGROUND TO THE INVENTION

The application of a Free Piston Stirling Engine (FPSE) with a LinearAlternator (LA) to generate electricity in Domestic Combined Heat andPower (DCHP) units that provide hot water and central heating in adomestic environment is well known. Referring to FIG. 1, there is showna typical power characteristic curve of a FPSE/LA, which is similar tothat shown in U.S. Pat. No. 4,642,547. The power output of a FPSE/LAtherefore depends on the head temperature. It may be understood that theamplitude and polarity of magnetic flux created by the permanent magnetschanges when the power piston moves relative to the armature winding(stator). Power modulation methods which are drawn in FIG. 1 may includeoperating at a constant voltage or at a constant output power.

As explained in U.S. Pat. No. 4,873,826, the output power of a FPSE/LAis a function of the engine heat exchanger temperature ratio (inaccordance with Carnot), operating frequency, mean pressure, andvolumetric displacement of the displacer and the power piston. Thisdocument and U.S. Pat. No. 6,871,495 propose power modulation techniquesbased on mechanical controls. These present several drawbacks such asslow transient response, low reliability and higher cost.

However, power modulation can be achieved in the electrical side bymeans of regulating the load connected to the FPSE/LA which indirectlycontrols the current flowing through the alternator. The mechanicalforce acting on the piston (F_(r)) is proportional to the currentflowing through the stator coil (i) and it provides an effectiveinertial load to the FPSE. This is shown with variation in time (t) inthe equation below, where a is the LA motor constant.

F _(p)(t)=α·i(t).

It is therefore desirable that such low-inertia generators can beprovided with a suitable impedance across the generator terminals,irrespective of load demand. An impedance sensed by the alternator thatis too high or too low could result in over-voltage, waveformdistortion, and in extreme cases (such as an open or short circuit)physical damage of the generator engine.

The engine or alternator are normally assured of being presented with areasonably stable impedance when connected directly to the electricalmains supply. However, there is often no inherent protection for theengine or alternator if they are used to provide electrical energy toconnected appliances when disconnected from electrical mains, as in thecase of a grid power blackout. A large number of techniques have beenproposed for regulating the voltage in such scenarios.

As explained in U.S. Pat. No. 6,856,107, the induced voltage may bedetermined by assuming a sinusoidal flux waveform and, according toFaraday's law, its peak value is proportional to the amplitude of thepower piston position. As can be seen from FIG. 1, the relationshipbetween generated power and voltage seems to follow a quadraticrelationship when the engine operates at a constant temperaturegradient. Power modulation is performed modifying the operation point ofthe FPSE/LA, by controlling the head temperature and the load connectedto the FPSE/LA.

Several power modulation control methods are known for keeping aconstant displacer-to-piston stroke ratio and their relative phase angleby controlling the load connected to the FPSE/LA. A simple controlstrategy is described in U.S. Pat. No. 4,873,826, in which the windingratio of an autotransformer is adjusted to maintain a constant voltageafter the tuning capacitor. This strategy may be suitable for grid andoff-grid applications. Another control method suitable for laboratorytesting is based on a variable frequency power supply (inverter at afixed output voltage), one autotransformer and a ballast load which isshown in U.S. Pat. No. 7,200,994.

Other approaches use an electronic load, which is a circuit thatexploits the electrical characteristics of a power electronics topologyin order to control the load impedance. Several known power electronictopologies may implement an electronic load. For example, U.S. Pat. No.6,871,495 describes connecting different resistive loads to achievevoltage regulation in a DC bus, which is the rectified FPSE/LA outputvoltage after a tuning capacitor. To overcome the disadvantages of usinga tuning capacitor to compensate the winding inductance of thealternator, an active rectifier has also been proposed in U.S. Pat. No.6,856,107, U.S. Pat. No. 7,453,241 and U.S. Pat. No. 6,871,495. Assuggested in U.S. Pat. No. 7,453,241 and U.S. Pat. No. 6,871,495, theactive rectifier bridge transistors may be switched to control the phaseof a SPWM (Sinusoidal Pulse Width Modulation) signal, until thealternator current is in phase with piston position or alternator EMF.The load is then connected in the DC bus and regulated using a voltagecontroller.

Several analogue and digital control techniques were proposed for theelectronic loads to achieve power modulation of a Stirling engine. U.S.Pat. No. 7,453,241 proposes a strategy based on a constant voltagecontrol in the DC bus side by means of using a hysteresis controller.US-2009/224738 and U.S. Pat. No. 6,871,495 consider digital controltechniques using a reference sine wave in synchronism (or phase) withthe FPSE/LA piston position or EMF.

Referring first to FIG. 2A, there is shown a first equivalent circuitfor an existing regulator technology, which seeks to ensure a stableimpedance in such cases using an electronic load. A generator comprisinga FPSE/LA 10 has a tuning capacitance 12. The output voltage of thegenerator is measured by voltmeter 14 and an electronic load 22 is alsoacross the generator output. The electronic load 22 is controlled by avoltage controller 20, which bases its control on the voltage measuredby voltmeter 14.

However under such conditions, the load actually corresponds to theconnected appliances that are connected across the electrical output ofthe alternator. These are termed customer loads and they may vary fromzero up to the full rated output of the alternator. Desirably, the powerof the engine should power up customer loads instead of damping theFPSE/LA generated power without any particular use.

Referring next to FIG. 2B, there is shown a second equivalent circuitfor an existing regulator technology, which has similar components tothose of FIG. 2A and these are identified by identical referencenumerals. Customer load 30 is also across the generator output. Whenappliances are first connected to the generator these loads may demand“inrush” currents that are greatly in excess of those normally providedby the alternator. Inductive loads could also require high levels ofcurrent for short periods of time. Therefore, an adequate regulationstrategy is demanded to ensure that such a low inertia generator ispresented with stable impedance under all load demand conditions.Therefore, typical power modulation techniques are based on controllingthe impedance connected to the FPSE/LA as it can provide control of thedisplacer to piston stroke ratio and their relative phase angle.

Referring now to FIG. 2C, there is shown a third equivalent circuit foran existing regulator technology, which has similar components to thoseof FIGS. 2A and 2B and these are identified by identical referencenumerals. In addition to the voltage control 20, a power control block40 is also placed at the generator output. An equivalent circuit for aregulator technology in accordance with US-2009/189589 (commonlyassigned with this invention) is shown in FIG. 2D. Again, where the samecomponents as those of FIG. 2C are used, they are identified byidentical reference numerals. The power control block 40 of FIG. 2Ccomprises a voltmeter 46 and a current meter 48, which providemeasurements to power control block 42. This controls an AC chopper 44to affect the output voltage across the customer loads 30.

Determining the control signals for the voltage control block 20 andpower control block 42 is not straightforward. US-2009/189589 suggests atechnique for determining an error signal that can be used to modulatethe AC input signal in order to obtain a regulated AC output signal.Referring now to FIG. 3A, there is shown a schematic diagram of a methodfor regulating the AC signal, as described in this document. The ACinput signal is sampled to produce a sampled AC signal as shown by inputwaveform 52. This is provided to full-wave rectifier 56 and therectified AC signal 58 results. In parallel, the sampled AC signal 52may be used to generate trigger pulses 62 to coincide with the sampledAC signal 52 crossing through zero volts. This zero crossing may bedetected using software and may use digital filtering to remove theeffects of noise around the zero crossing and make use of softwarepattern matching to improve face synchronisation. A computer 64 usesthese trigger pulses 62 to generate a synchronised reference signal 66.The reference signal 66 corresponds to a sinusoid, but with onlypositively extending lobes such that it is equivalent to a full-waverectified AC signal.

In ratio-metric comparison block 70, the scaled AC signal 58 issubtracted from the reference signal 66 to produce an error signal 72.In other words, instantaneous values are subtracted from instantaneousvalues. To ensure that only positive values are obtained, an offset isintroduced. For example, this subtraction may be implemented in adifference amplifier operating with a suitable offset.

This error signal 72 is not only a function of the amplitude differencebetween the scaled AC signal 58 and the reference signal 66, it is alsoa function of the phase of the AC input signal. This phase variation maybe removed at 74 by a multiplier chip 76 that operates to divide theerror signal by the reference signal 66 to provide a percentage errorsignal 78. This percentage error signal 78 may then be used to modulatethe AC input signal.

This is a relatively fast-response control loop, as instantaneouschanges in the error signal are immediately reflected in changes to themodulation. Referring now to FIG. 3B, there is shown a further schematicdiagram, illustrating the generation of the exact error signal forcontrolling the modulation. The AC input voltage 52 and the designoutput voltage 66 are provided to the ratio-metric comparison block 70,which generates the percentage error signal 72. This is passed to aproportional controller 76 which generates the analogue error signal 78.This is compared with the output of a ramp generator 77 using acomparator 79 to generate a Pulse Width Modulation (PWM) signal 81,which is fed to an electronic load such as an AC/AC Buck regulator (notshown), also called an AC/AC chopper.

This control strategy and error definition effect power control (thatis, regulation of the voltage when it begins to drop, indicating that ahigher than normal load is being applied), because the error signal ispositive when the generator output (as scaled) is less than thereference signal. This may deal with in-rush currents, as explainedabove. The error definition changes when voltage control is required(that is, regulation of the voltage when the load applied is within anormal range), such that the error signal is positive when the generatoroutput (as scaled) is greater than the reference signal.

By generating an error signal that is expressed as a fraction (that is,a percentage) of the reference signal, the error signal magnitude isessentially independent from the LFPSE/LA AC voltage level. This makesit suitable for use with a fast proportional controller, such asproportional controller 76. This complex control strategy appears to bewell-suited to generators based on low inertia engines, for example,Stirling engines, in which a regulation strategy that minimises the riskof mechanical failure is demanded. However, it requires a significantamount of regulation-specific processing.

SUMMARY OF THE INVENTION

Against this background, the present invention provides a regulator forthe electrical output of a generator, comprising: an input for receivinga signal indicative of at least one characteristic of the electricaloutput; a first, relatively fast-response sub-controller configured toprovide a first control signal on the basis of the at least onecharacteristic; a second, relatively slow-response sub-controllerconfigured to provide a second control signal on the basis of the atleast one characteristic; and an output, arranged to provide a combinedcontrol signal to adjust the electrical output based on the first andsecond control signals.

This approach regulates the generator output using two simple controlloops: a fast-response controller; and a slow-response controller.Whilst a complex fast-response controller relying on instantaneous ornear-instantaneous measurement has been used in existing systems, byreplacing this with two simpler controllers, the overall regulator isless complex, costly, power consuming and easier to manage. Aslow-response controller can be based on statistics of the input, suchas an average, particularly a time-averaged input.

Preferably, the generator comprises a Stirling engine. In the preferredembodiment, the generator comprises a Free Piston Stirling Engine (FPSE)or a Linear Free Piston Stirling Engine (LFPSE). This may be usedtogether with a Linear Alternator (LA). The combination may be referredto as FPSE/LA or LFPSE/LA. Generators comprising another type of lowinertia engine may also be considered, such as thermoacoustic engines.

The first and second sub-controllers are typically of different types.The first sub-controller is preferably a feedforward controller and morepreferably comprises a clamp controller. Additionally (oralternatively), the second sub-controller may comprise an errorcompensator, as is known for a typical Single Input Single Output (SISO)control strategy. A range of types of error compensator can be used. Thepreferred type is a form of Proportional-Integral-Derivative (PID)controller. This may be a Proportional controller or aProportional-Integral (PI) controller in preferred embodiments. This maybe easier to tune and it may work well when the plant (that is, thegenerator) characteristic is unknown. Other types of error compensatormay include lead and lag compensators, as long as the closed systemresponse is stable.

Advantageously, the second sub-controller uses a DC level as a reference(particularly when the second sub-controller comprises a PID-type ofcontroller). This assists in making the regulator less complex and proneto problems, because it is not necessary to synchronise the referencevoltage with the generator output when a DC level is used. Moreover, itis easier to generate a stable DC level. Additionally or alternatively,the second sub-controller beneficially compares a time-averaged valuefor the at least one characteristic of the electrical output with areference. This provides a simple and efficient slow-responsesub-controller. Moreover, the controller can also cope withnon-sinusoidal waveforms as well as sinusoidal ones.

The at least one characteristic may comprise one or more of the:voltage; current; and power of the electrical output. In preferredembodiments, only the voltage is used.

In one embodiment, the first and second sub-controllers are configuredfor a voltage control mode. The voltage control mode may regulate thevoltage on the basis that the load impedance (or resistance in the usualcase where no reactive loads are applied) is greater than a rated value.

In another embodiment, the first and second sub-controllers areconfigured for a current control mode. Contrastingly, the currentcontrol mode may regulate the voltage on the basis that the loadimpedance (or again, resistance) may be lower than the rated level and,as a consequence, excess power is being generated that needs to besafely dissipated. If the generator is overloaded, this generally meansthat the load impedance connected to it is lower than a desired value (a“desired impedance”). The lower the value of the impedance connected tothe generator, the greater is the current for a given voltage (in otherwords, the greater the output power). The current control mode may beused when there is a high demand of current, which is seen as the loadimpedance being lower than the desired impedance. The voltage controlmode may address the situation when the load impedance increases atconstant power output, thereby causing the voltage to increase.

Preferably, the first sub-controller is configured to provide a firstcurrent-control signal and a first voltage-control signal. Thus, thefirst control signal may comprise a first current-control signal and afirst voltage-control signal. Then, the second sub-controller may beconfigured to provide a second current-control signal and a secondvoltage-control signal. Thus, the second control signal comprises asecond current-control signal and a second voltage-control signal. Here,the output may be arranged to provide the combined control signalcomprising: a combined current-control signal to adjust the electricaloutput based on the first and second current-control signals and acombined voltage-control signal to adjust the electrical output based onthe first and second voltage-control signals.

In the preferred embodiment, the first sub-controller comprises: a firstcurrent-control sub-controller configured to provide the firstcurrent-control signal on the basis of the at least one characteristic;and a first voltage-control sub-controller, configured to provide afirst voltage-control signal on the basis of the at least onecharacteristic. Then, the second sub-controller may comprise: a secondcurrent-control sub-controller, configured to provide the secondcurrent-control signal on the basis of the at least one characteristic;and a second voltage-control sub-controller, configured to provide thesecond voltage-control signal on the basis of the at least onecharacteristic.

Each of the first and second sub-controllers may comprise a respectiveerror signal generator. Each error signal generator may be configured togenerate a respective error signal by comparing the at least onecharacteristic with a respective reference value. Also, each errorsignal generator may be further configured to generate the respectivecontrol signal so as to minimise the respective error signal. The firstand second sub-controllers may be configured to generate the respectiveerror signal by determining the difference between the at least onecharacteristic and the respective reference value. Each of the first andsecond current-control sub-controllers may generate the respective errorsignal by determining a difference between the at least onecharacteristic and the respective reference value. If the at least onecharacteristic comprises voltage or displacement, the first and secondcurrent-control sub-controllers may generate the respective error signalby subtracting the at least one characteristic from the respectivereference value. If the at least one characteristic comprises current orpower, the first and second current-control sub-controllers may generatethe respective error signal by subtracting the respective referencevalue from the at least one characteristic.

Advantageously, the combined control signal comprises at least one PulseWidth Modulation, PWM, signal. In the preferred embodiment, the combinedcontrol signal comprises: the combined current-control signal; and thecombined voltage-control signal, each of which comprises a PWM signal.Advantageously, the first and second control signals are PWM signals(and likewise, the first and second current-control signals and thefirst and second voltage-control signals). The combined control signal(or signals) may then be relatively simple to generate by applying alogical OR operator to the respective first and second control signals.

In some embodiments, the input comprises terminals configured to receivethe electrical output from the electrical generator. Then, the regulatormay further comprise an electronic load, arranged across the inputterminals and configured to receive the combined control signal(preferably, the combined voltage-control signal) from the output and toset its resistance based on the received combined control signal, toadjust the electrical output thereby. Electronic loads may provide afast, reliable and cost effective way to control impedance. Theelectronic load may take the form of a DC chopper (for instance,comprising a power electronics converter, such as a buck, boost orflyback topology, coupled to a fixed load) or some other form ofelectronically controlled resistance. This may be located after a tuningcapacitor at the output of the generator. Advantageously, the electronicload is responsive to the combined voltage-control signal.Alternatively, it may be deactivated and basically provide an opencircuit (effectively infinite resistance).

Preferably additionally (although possibly alternatively), the regulatormay further comprise an AC converter, arranged at the input terminalsand configured to receive the combined control signal from the outputand to adjust the electrical output in accordance with the receivedcombined control signal. Again, the AC converter is beneficiallyresponsive to the combined current-control signal. Otherwise, it may bedeactivated. The output (or load) may be automatically disconnected (orde-energised). The AC converter may be an AC/AC converter, such as an ACchopper (AC/AC buck, boost or full bridge, for example) or an AC/DCconverter.

The input is beneficially configured to receive the electrical outputfrom the electrical generator. Then, the input may comprise a signalprocessing module arranged to generate a second signal indicative of theat least one characteristic by processing of the electrical output. Thefirst and second sub-controllers are advantageously responsive to thesecond signal indicative of the at least one characteristic. Optionally,the signal processing module is configured to generate the second signalindicative of the at least one characteristic as a scaled version of theelectrical output. Advantageously, the signal processing module isconfigured to generate the second signal indicative of the at least onecharacteristic as a rectified version of the electrical output.

In a second aspect, there is provided a method of regulating theelectrical output of a generator, comprising: receiving a signalindicative of at least one characteristic of the electrical output;generating a first control signal on the basis of the at least onecharacteristic using a first, relatively fast-response sub-controller;generating a second control signal on the basis of the at least onecharacteristic using a second, relatively slow-response sub-controller;and adjusting the electrical output using a combined control signal thatis based on the first and second control signals.

It will be understood that this method may comprise optional methodsteps corresponding with any one or more of the apparatus featuresdefined herein. It will also be appreciated that the present inventionmay be found in programmable logic (such as a Complex programmable logicdevice, CPLD, Digital Signal Processor, DSP, or Field Programmable GateArray, FPGA) configured to carry out any method as disclosed herein or acomputer program, configured when operated on a processor to carry outany method as disclosed herein.

The combination of any of the apparatus or method features describedherein or both is also provided even if not explicitly disclosed.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention may be put into practice in various ways, one of whichwill now be described by way of example only and with reference to theaccompanying drawings in which:

FIG. 1 shows a power characteristic of a generator based on a FreePiston Stirling Engine with a Linear Alternator (FPSE/LA);

FIGS. 2A to 2D depict equivalent circuits for existing regulatortechnologies;

FIG. 2E illustrates an equivalent circuit for a regulator technology inaccordance with the invention;

FIGS. 3A and 3B show schematic diagrams for a known approach togenerating an error signal for regulation of the output voltage for agenerator based on a FPSE/LA;

FIGS. 4A and 4B show schematic diagrams for an approach to generating anerror signal for regulation of the output voltage for a generator basedon a FPSE/LA in accordance with the present invention;

FIGS. 5A and 5B illustrate example voltage waveforms as part of thegeneration of an error signal in accordance with the embodiment of FIGS.4A and 4B;

FIGS. 6A and 6B depict example voltage waveforms and duty cyclecharacteristics for the operation of a clamp controller in accordancewith the embodiment of FIGS. 4A and 4B respectively;

FIG. 7 shows how thresholds for controlling the operation of theembodiment of FIGS. 4A and 4B can be set for stable operation;

FIG. 8 shows how thresholds for controlling the operation of theembodiment of FIGS. 4A and 4B can be set where stable operation may notalways be possible;

FIG. 9 shows how thresholds for controlling the operation of theembodiment of FIGS. 4A and 4B can be set for unstable operation;

FIG. 10 shows how thresholds for controlling the operation of theembodiment of FIGS. 4A and 4B can be set such that stable operation isnever possible;

FIG. 11 depicts a circuit and equivalent circuits for scaling thevoltage for use in the embodiment of FIGS. 4A and 4B;

FIG. 12 shows a circuit for a Proportional-Integral controller for usein the embodiment of FIGS. 4A and 4B;

FIG. 13 shows a circuit for a clamp controller for use in the embodimentof FIGS. 4A and 4B;

FIG. 14 shows a circuit for a ramp generator for use in the embodimentof FIGS. 4A and 4B;

FIG. 15 illustrates a generalised electronic load for use in theembodiment of FIGS. 4A and 4B; and

FIG. 16 shows a non-preferred design of electronic load.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

Referring first to FIG. 2E, there is shown an equivalent circuit for aregulator technology in accordance with the invention. Some of thecomponents of this circuit are the same as FIG. 2D and these areindicated by identical reference numerals. A power control block 43 usesthe voltage measured by voltmeter 14 to generate a control signal forthe AC chopper 44. Thus, a current meter and a second voltmeter are notneeded. This circuit is intended to form a bridge between the alternatorof a DCHP unit (of which the LFPSE 10 forms a part), an electrical gridand also local electrical appliances to ensure that the signal producedby the alternator is suitable for injection into the grid, supply to theconnected appliances or both.

Control Approach

Referring next to FIG. 4A, there is shown a schematic diagram for anapproach to generating an error signal for regulation of the outputvoltage for a generator based on a FPSE/LA. This is embodied as avoltage control block 100. Where the features of this approach are thesame as other drawings, these are indicated by identical referencenumerals. Two controllers act in parallel depending on the FPSE/LAoutput voltage 52. These are a clamp controller 120 and an errorcompensator 130. The block diagram also comprises: a scaling andrectification block 110; a triangle or ramp signal generator 142; afirst comparator 140; a second comparator 145; and a logical OR gate147.

An output of a precision rectifier in the scaling and rectificationblock 110 is provided to an overvoltage protection block (not shown).The output of the overvoltage protection block provides an interface toan external shutdown circuit (as described below). The scaling andrectification block 110 also provides an input to the clamp controller120 and the error compensator 130.

The clamp controller 120 provides a relatively fast acting response, toensure there is load present on the engine when the voltage of theFPSE/LA is larger than a certain threshold. The clamp controller 120provides fast transient changes in the load connected to the generator.

The error compensator 130 provides a relatively slow acting responsecontroller. It is intended to achieve long term load regulation. Theerror compensator is based on a Proportional-Integral (PI) controller.

Both the clamp controller 120 and the error compensator 130 receive thesame input, which a scaled and rectified version of the FPSE/LA ACoutput from the scaling and rectification block 110. The errorcompensator 130 acts in part as a comparator, comparing the scaled andrectified version of the FPSE/LA AC output with a DC voltage reference135. The DC voltage reference 135 represents the desired FPSE/LA outputvoltage.

The clamp controller 120 and the error compensator 130 each provide arespective output. These are two control signals for adjusting theoutput voltage seen by customer loads. Each of these two signals is thenprovided to a respective comparator 140, 145. The other input to thefirst comparator 140 and the second comparator 145 is commonly providedby the output of the triangle or ramp signal generator 142. Thus, theoutput of each of the first comparator 140 and the second comparator 145is a Pulse Width Modulation (PWM) signal. These PWM control signals arecombined using logical OR gate 147 to provide a single PWM voltagecontrol signal 148, which is provided to the electronic load 22. Theelectronic load 22 can be used to dissipate power as heat and may formpart of a heating system, such as an immersion heater for generating hotwater.

A circuit in accordance with this block diagram is implemented usingrelatively low cost analogue electronics and it does not require anymicrocontroller or expensive analogue multipliers. It also provides goodoutput voltage regulation with relatively few components and as a singlesupply circuit.

The clamp controller 120 and error compensator 130 are designed for avoltage control mode, in which the voltage output from the FPSE/LA is nolower than a desired level. In this case, the impedance seen by thelinear alternator can be decreased in order to dissipate excess powergenerated by the FPSE/LA and avoid the voltage seen at the customer loadincreasing. However, it does not deal with high inrush currents due totransient or reactive loads.

In order to overcome this problem, another extra control system may beprovided to check that:

-   -   the power demanded from the customer loads does not exceed the        maximum power available for a given FPSE/LA voltage;    -   the current demanded from the alternator is not going to be        larger than the saturation current which can result in damage to        the engine as the piston over-travels;    -   the inrush currents can be provided at lower output voltage        levels.

This new control system may act as a constant power source. Referringnext to FIG. 4B, there is shown a schematic diagram for an approach togenerating an error signal for regulation of the output voltage for agenerator based on a FPSE/LA for power control. This is embodied as apower control block 150. Where the features of this approach are thesame as other drawings, these are indicated by identical referencenumerals. As with the voltage control block 100, a clamp controller 160and an error compensator 130 act in parallel depending on the FPSE/LAoutput voltage 52. This operates in a similar fashion to the voltagecontrol block 130. However, the error compensator 130 compares thescaled and rectified version of the FPSE/LA AC output with a DC voltagereference 170, that may be different from the DC voltage reference 135used by the voltage control block 100. Also, the clamp controller 160and an error compensator 130 use a different error calculation from thevoltage control block 100. This will be discussed below. A single PWMpower control signal 149 is provided to the AC chopper 44.

The power control method with a voltage control strategy as shown inFIG. 4B can supply inrush current for short periods of time. When theFPSE/LA voltage after the tuning capacitor 12 is lower than the desiredoutput voltage the FPSE/LA is considered to be overloaded. Thiscondition may occur with loads demanding inrush currents for a certainperiod of time. Under an overload condition the maximum power availableof the generator will decrease in a quadratic way until the generatorstops. This drawback has proved to be overcome by stepping the generatorvoltage down using an AC/AC Buck converter to deliver higher currents atlower voltages by keeping the maximum power demanded by the load belowthe maximum power that the FPSE/LA can supply at a constant outputvoltage regulation.

A Grid Independent Module (GIM) unit is based on both the voltagecontrol block 100 and the power control block 150. The voltage controlblock 100 achieves power modulation of the FPSE/LA and the power controlblock 150 acts as a constant power source to deal with inrush currents.For the power control mode, the PWM control signal 149 controls theAC/AC Buck regulator 44. This affects the voltage seen by customer loads30, allowing a higher current to be drawn. For the voltage control mode,the PWM control signal 148 controls the electronic load 22. This maycause the voltage output of the linear alternator 210 to be reduced, ifthe impedance of the electronic load 22 is reduced. The implemented PWMoutput signal 148 of the voltage controller can be enabled or disabledexternally.

The power control mode seeks, at least in part, to mitigate the problemof inrush currents. The effect of inrush currents in the dynamics of theFPSE/LA can be explained from a mechanical viewpoint. The force actingon the piston depends on the pressure wave generated due to the Stirlingcycle. A steady-state balance between the force acting on the piston andthe opposing force depending on the alternator current is reached undernormal conditions. The alternator from a mechanical/dynamic point ofview behaves like a spring mass damper system.

To understand the effect of inrush currents the equivalent mechanicalsystem explaining the behaviour of the linear alternator is simplifiedto a spring. The spring constant of the alternator is going to bedependent on the alternator saturation condition. When a current islarger than the maximum current that saturates the alternator, thespring constant is going to decrease dramatically and the piston canover-travel (the force opposing the piston movement may be much lowerdepending on displacement). It should be noted the engine has somespring magnets that can provide some degree of protection to overloads,but large inrush currents remains a problem with this type of generator.

A more detailed description of the operation of the controllers is nowprovided.

Scaling and Rectification Block

Referring to FIG. 5A, there are shown example voltage waveforms as partof the generation of an error signal, specifically an example for thewaveform generated by the scaling and rectification block 110. TheFPSE/LA voltage 300 is an AC sinusoidal voltage (for example, 240 Vrms).This voltage is scaled to DC levels (from 0 to 5V with a 2.5V DCoffset).

Referring to FIG. 5B, there is shown an example voltage waveform as partof the generation of an error signal, specifically an example for thewaveform generated by the precision rectifier. The scaled AC signal 310is rectified to rectified signal 320, so that an error voltage can bedefined. The average value, V_(avg), of the rectified signal 320 isproportional to the FPSE/LA voltage 300.

Referring now to FIG. 11, there is depicted a circuit and equivalentcircuits for scaling the voltage within the range from 0 to 5V. Thegoverning equations for this circuit are as follows.

$V_{mac} = {{V_{ac}\frac{\left( {R_{2}\text{/}\text{/}Z_{C}\text{//}R_{1}} \right)}{R_{3} + \left( {R_{2}\text{/}\text{/}Z_{C}\text{/}\text{/}R_{1}} \right)}} = {V_{ac}\frac{R_{2}Z_{C}R_{1}}{{R_{3}\left( {{R_{1}Z_{C}} + {R_{2}R_{1}} + {R_{2}Z_{C}}} \right)} + {R_{2}Z_{C}R_{1}}}}}$$V_{mdc} = {{V_{dc}\frac{\left( {R_{3}\text{/}\text{/}R_{2}\text{//}Z_{C}} \right)}{R_{1} + \left( {R_{3}\text{/}\text{/}R_{2}\text{/}\text{/}Z_{C}} \right)}} = {V_{dc}\frac{R_{3}R_{2}Z_{C}}{{R_{1}\left( {{R_{2}Z_{C}} + {R_{3}Z_{C}} + {R_{3}R_{2}}} \right)} + {R_{3}R_{2}Z_{C}}}}}$

Error Compensator

In voltage control mode, the error compensator 130 receives a DC voltagereference within 2.5 to 5V, representing the desired output voltage.This is shown as V_(ref) in FIG. 5B. The error is defined as the FPSE/ACscaled and rectified voltage 320 minus the DC voltage reference, that isV_(avg)−V_(ref). The PI error compensator 130 provides an output signalthat minimizes this error.

In power control mode, a DC voltage reference within 0 to 2.5Vrepresents the desired output voltage. The error is defined as thedesired FPSE/LA AC output voltage minus the FPSE/AC scaled and rectifiedvoltage 320, that is V_(ref)−V_(avg). The error compensator 130 againprovides an output signal that minimizes this error.

Analogue or digital error compensator techniques can be used. Ananalogue PI controller is currently used as it provides a good transientresponse, simplicity and low cost. Other analogue compensationstrategies can be implemented, such as suggested in Chetty, P. R. K.,“Modelling and design of switching regulators”, IEEE Transactions onAerospace and Electronic Systems, 1982, AES-18 (3), p. 333-344.

Referring then to FIG. 12, there is shown a circuit for an analogueProportional-Integral controller. The frequency response of this circuitis given by the following equation.

${V_{0}(s)} = {{V_{ref}(s)} + {\frac{\left( {R_{2} + \frac{1}{sC}} \right)}{R_{1}}\left( {{V_{ref}(s)} - {V_{1}(s)}} \right)}}$

The time domain response of the circuit is given by the followingequation.

${V_{0}(t)} = {V_{ref} + \overset{\overset{{proportional}\mspace{14mu} {action}}{}}{\frac{R_{2}}{R_{1}}\overset{\overset{{error}\mspace{14mu} {signal}}{}}{\left( {V_{ref} - {V_{1}(t)}} \right)}} + \overset{\overset{{Integral}\mspace{14mu} {action}}{}}{\int_{0}^{t}{\frac{\overset{\overset{{error}\mspace{14mu} {signal}}{}}{\left( {V_{ref} - {V_{1}(t)}} \right)}}{R_{1} \cdot C}\ {t}}}}$

Clamp Controller for Voltage Control Mode

As the FPSE/LA is a low inertia generator, the clamp controller 160(feed forward controller) is implemented to provide fast transientprotection, especially if the engine voltage is higher than 260 Vrms.This can result in irreparable damage due to an overstroke. The clampcontroller 160 only acts when the rectified engine voltage is higherthan a certain value (for instance, 240 Vrms). This circuit ensuresthere is always a load present when the engine voltage is higher than acertain clamp threshold (V_(clamp) _(_) _(th), a predefined value isassociated with 240 Vrms, for example). The clamp circuit is highlydesirable when there are large step variations in the load connected tothe generator as it provides protection to transient loads.

The clamp circuit provides a real time response. When the scaled andrectified FPSE/LA AC signal is greater than the clamp threshold, theclamp controller provides an output voltage proportional to the voltagedifference. The error for the clamp circuit for voltage control mode isdefined as:

${error}_{clamp} = \left\{ {\begin{matrix}\; & {{{scaled}\mspace{14mu} {FPSE}\text{/}{LA}_{{ac}\mspace{14mu} {signal}}} - V_{{clamp}\_ {th}}} \\0 & {{{if}\mspace{14mu} V_{{clamp}\_ {th}}} > {{scaled}\mspace{14mu} {FPSE}\text{/}{LA}_{{ac}\mspace{14mu} {signal}}}}\end{matrix}.} \right.$

Referring next to FIG. 6A, there are shown example voltage waveforms andduty cycle characteristics for the operation of the clamp controller involtage control mode. The duty cycle signal for the voltage control isvalid for an electronic load based on a DC Buck converter. If otherelectronic load implementations based on different power electronictopology may require different duty cycle signal versus alternatorvoltage characteristics. The voltage threshold and the gain can beadjusted with a pair of potentiometers.

The rectified voltage is also used for the overvoltage protection block.An overvoltage event could occur as an example under the unlikely eventof an electronic load failure. If the engine voltage is higher than,say, 260 Vrms a solid state switch or relay is triggered for apredefined period of time by a monostable. As soon as an overvoltagetrip appears, the LFPSE/LA is shut down.

The duty cycle for the PWM signal can be defined for a voltage higherthan the clamp threshold. A possible clamp transfer response for voltagecontrol mode is shown in FIG. 6A.

Referring to FIG. 6B, there are shown example voltage waveforms and dutycycle characteristics for the operation of the clamp controller in powercontrol mode. The duty cycle signal for the power/current control isvalid for the AC/AC Buck converter, although again other electronic loadimplementations based on different power electronic topology may requiredifferent duty cycle signal versus alternator voltage characteristics.The clamp controller is modified to use an inverted version of therectified voltage. When the scaled and rectified FPSE/LA AC signal islower than the clamp threshold for current control (which will be lessthan the clamp threshold for voltage control), the clamp provides anoutput voltage proportional to the voltage difference.

${error}_{clamp} = \left\{ {\begin{matrix}\; & {V_{{clamp}\_ {th}} - {{scaled}\mspace{14mu} {FPSE}\text{/}{LA}_{{ac}\mspace{14mu} {signal}}}} \\0 & {{{if}\mspace{14mu} {scaled}\mspace{14mu} {FPSE}\text{/}{LA}_{{ac}\mspace{14mu} {signal}}} > V_{{clamp}\_ {th}}}\end{matrix}.} \right.$

The error is defined as the FPSE/AC output voltage scaled, rectified andinverted voltage minus the desired FPSE/LA AC output voltage(represented by a DC voltage value).

Referring next to FIG. 13, there is shown a circuit for a clampcontroller. Analysis of this linear circuit yields the followingequations.

$v^{+} = {{\frac{\left( {V_{1} - X_{{ref}_{1}}} \right)}{\left( {R_{1} + R_{2}} \right)}R_{2}} + {X\; {ref}_{1}}}$$V_{01} = {{\left( {1 + \frac{R_{2}}{R_{1}}} \right)v^{+}} + {V_{{ref}\_ {clamp}}\left( {- \frac{R_{2}}{R_{1}}} \right)}}$$V_{02} = {{\left( {1 + \frac{R_{3}}{R_{4}}} \right)V_{01}} + {X_{{ref}\; 1}\left( {- \frac{R_{3}}{R_{4}}} \right)}}$

Effect of Voltage and Power Control Modes

As shown in FIG. 2E, a voltage control block 20 may achieve powermodulation of the engine and a power control block 43 may cope withloads that demand high inrush currents. These two controllers operatewith different error definitions, as explained above. For the voltagecontrol the error is positive when the FPSE/LA voltage is higher thanthe desired reference voltage. Therefore, a higher load must beconnected to the engine to minimize the error and achieve FPSE/LA outputvoltage regulation. For the power (current control) the error ispositive when the FPSE/LA voltage is lower than the desired voltage.

As a result, there are two voltage settings for each controller,identifying a voltage on or above which voltage control mode is enabled(Vref_VC) and a voltage on or below which power control mode is enabled(Vref_PC). Referring first to FIG. 7, there is shown how thresholds forcontrolling the operation of the controller can be set for stableoperation. The reference voltage for the Voltage Control is at least fewvolts higher than the reference voltage for the Current/Power Control.Interactions between the Voltage Control (VC) and the Power control (PC)are avoided thereby.

The two different reference voltages implemented in this way providehysteresis and avoid hunting (that is repeated alternation between themodes resulting from noise in the signal causing repeating crossings ofthe threshold). This is discussed in more detail in US-2009/224738. Forexample, when the FPSE/LA is operating in voltage control mode with areference voltage of 220 Vrms, a voltage drop to 220 Vrms or below islarge enough to indicate excessive current demand to cause the switch topower control mode. When the power control mode is dominant, a rise to225 Vrms may be used to indicate the current demand is normal to causethe voltage controller to dominate instead.

Referring then to FIG. 8, there is shown how thresholds for controllingthe operation of the controller can be set where stable operation maynot always be possible. Theoretically, if the voltage thresholds for theVC and PC are set to the same value (or very close values), thecontroller should operate without a problem. However, noise present inthe FPSE/LA in a real scenario could lead to undesirable interactionbetween both the VC and PC algorithms. Hence, the controller couldbecome unstable.

Referring now to FIG. 9, there is shown how thresholds for controllingthe operation of the controller can be set for unstable operation.Hunting is likely to occur in this case. Referring finally to FIG. 10,there is shown how thresholds for controlling the operation of thecontroller can be set such that stable operation is never possible. Thepower (current) control mode is always active, stepping down the voltagein the customer loads.

Ramp Generator

Referring now to FIG. 14, there is shown a circuit for a ramp generator.The ramp generator is implemented using two comparators 401, 402 and oneRS flip flop 403. The oscillation period, T, follows the followingequations.

T = T_(charge) + T_(discharge), where$T_{discharge} = {{\tau \cdot {\ln \left( \frac{X_{{ref}\; 1}}{X_{{ref}\; 2}} \right)}}\mspace{14mu} {and}}$$T_{charge} = {\tau \cdot {{\ln \left( \frac{X_{{ref}\; 2} - V_{s}}{X_{{ref}\; 1} - V_{s}} \right)}.}}$

Electronic Load

An electronic load is a variable load that exploits the electricalcharacteristics of a power electronic topology. This is used forvoltage-control.

Referring to FIG. 15, there is illustrated a generalised electronicload, comprising a power electronics converter coupled to a fixed load.Depending on the type of power electronic topology and mode of operationof the electronic load, the Gain function (G) may be different and theresultant impedance relationship will vary accordingly. For example, abuck converter operating in CCM (Continuous Conduction Mode) willreceive a PWM signal with a duty cycle defined as d, which is a variablewithin the range [0,1]. Then, its Gain will be defined as

$G = {\frac{V_{out}}{V_{in}} = {d.}}$

For a boost converter in CCM, the Gain will be defined as

$G = {\frac{V_{out}}{V_{in}} = {\frac{1}{1 - d}.}}$

For an isolated converter such as a flyback in CCM, the gain is afunction of the duty cycle and the transformer windings ratio, n.

$G = {\frac{V_{out}}{V_{in}} = {{\frac{1}{1 - d} \cdot n} = {{f\left( {d,n} \right)}.}}}$

For other converters, the Gain functions can also be determined.

In general, it can therefore be seen that the output voltage of theelectronic load is a function of the duty cycle, as given by thefollowing relationship:

V _(out) =V _(in) ·G=V _(in) ·f(d).

The following relationships are well known.

$P_{in} = {{V_{in} \cdot I_{in}} = {{\frac{V_{in}^{2}}{R_{in}}\mspace{14mu} P_{out}} = {{V_{out} \cdot I_{out}} = \frac{V_{out}^{2}}{R_{out}}}}}$

Using these and assuming a lossless power transformation (P_(out)=P_(in)or equivalently, η=P_(out)/P_(in)≈1), the relationship between the inputand output impedance can also be determined. Usually, the convertershows a performance less than 1 but we can assume it is one.

$P_{in} = {\frac{V_{in}^{2}}{R_{in}} = {P_{out} = \frac{\left( {V_{in} \cdot {f(d)}} \right)^{2}}{R_{out}}}}$$\frac{V_{in}^{2}}{R_{in}} = \frac{V_{in}^{2} \cdot {f^{2}(d)}}{R_{out}}$$R_{in} = \frac{R_{out}}{f^{2}(d)}$

Then, specific relationships can be identified for a specific electronicload with a specific f(d). For a buck converter in CCM:

f(d) = d  and$R_{in} = {\frac{R_{out}}{f^{2}(d)} = {\frac{R_{out}}{d^{2}}.}}$

For a boost converter in CCM:

${f(d)} = {\frac{1}{1 - d}\mspace{14mu} {and}}$$R_{in} = {\frac{R_{out}}{f^{2}(d)} = {{R_{out}\left( {1 - d} \right)}^{2}.}}$

For a flyback in CCM:

$R_{in} = {\frac{R_{out}}{f^{2}\left( {d,n} \right)} = {R_{out}\frac{\left( {1 - d} \right)^{2}}{n^{2} \cdot d^{2}}}}$

As can be seen, the input impedance for a flyback converter (withgalvanic isolation) is a function of the transformer winding ratio andthe duty cycle.

An electronic load can therefore be implemented with any type of powerelectronic topology. However, the buck converter has advantages in termsof simplicity and cost. If any reason (such as safety regulations)dictates that galvanic isolation is required for the fixed load (whichmay be an immersion heater), then another type of converter such as ahalf-bridge or push-pull could be used instead of the buck converter.

Another form of electronic load may use a variable autotransformer, withits wiper being moved to step up or down the voltage connected to thefixed load. Referring next to FIG. 16, there is shown an alternativetype of electronic load to a DC chopper and variable autotransformer. Adiscrete number of resistors (R₁, R₂, . . . , R_(n)) are provided inparallel, with each resistor being coupled to a respective switch (SW₁,SW₂, . . . , SW_(n)) that controls whether current passes through theresistor. These may be relays or solid state switches. A control blockprovides a control signal (or signals) to operate or deactivate theswitches accordingly. However, this approach is bulky, costly andgenerally cannot achieve fine voltage regulation (as only discrete stepsare possible).

AC Converter

For the power control (current control) of AC loads, any AC convertermay be used. If the load is supplied with AC, an AC/AC converter isused. This may be based on bidirectional switches or high frequencylinks. The type of AC/AC converter used may depend on the type of load.

The principle of operation is similar to that of the electronics load.The input impedance can be controlled by using a control signal (with aduty cycle). For the case of AC loads such as the GIM unit, the simplestAC/AC converter or power electronic topology to implement is a buckconverter. The AC/AC buck converter in continuous conduction mode (CCM)shows a gain proportional to the duty cycle. The voltages however arenormally sinusoidal though, compared with the traditional DC buck.According to the gain function (shown below), the input impedance can becontrolled by changing the duty cycle:

$G = {\frac{V_{out}}{V_{in}} = {d.}}$

Thus, the impedance relationship is also controlled by the duty cycle.

$R_{in} = {\frac{R_{out}}{f^{2}(d)} = \frac{R_{out}}{d^{2}}}$

However, there are many types of power electronics topology for ACconverters, which may include: AC/AC boost; AC/AC buck boost; and AC/ACwith isolation topologies (full bridge). For example, where the AC loadrequires galvanic isolation, an AC/AC full bridge is desirably used.

Essentially, the same concepts as applied to the electronic load canalso be applied to the AC converter. Although any power electronictopology may be suitable, economic constraints, reliability constraints(number of switches, stresses in the semiconductors) and other issuesmay limit the selection of AC converter topology. In the embodiment ofFIG. 4B, the AC/AC buck was selected in view of these issues.

ALTERNATIVES

Whilst a specific embodiment has been described, the skilled person maycontemplate various modifications and substitutions. For example, slowresponse and fast response controllers may be used. Even if the fastresponse controller is a clamp controller, the slow response controllerneed not be a PI controller, as a proportional controller or a PIDcontroller can be used, for instance.

Both voltage-control and current-control are provided in the preferredembodiment. Nevertheless, it will be understood that alternatives may bepossible. In particular, other implementations may providevoltage-control only or current-control only and it may be possible toimplement each or both selectively.

Whilst PWM control signals are used in the embodiment described herein(with some advantages), other types of signal may be used instead. Thecircuits for the components of the voltage control block and currentcontrol block can differ from those described above and the commoncomponents of the voltage control block and current control block may bedifferent from those suggested herein, for practical or efficiencyreasons. Analogue circuitry, digital circuitry and a combination of thetwo can be used to implement the invention. Programmable logic, firmwareor software can be used in addition or as an alternative.

1. A regulator for the electrical output of a generator, comprising: aninput for receiving a signal indicative of at least one characteristicof the electrical output; a first, relatively fast-responsesub-controller configured to provide a first control signal on the basisof the at least one characteristic; a second, relatively slow-responsesub-controller configured to provide a second control signal on thebasis of the at least one characteristic; and an output, arranged toprovide a combined control signal to adjust the electrical output basedon the first and second control signals.
 2. The regulator of claim 1,wherein the generator comprises a Stirling engine.
 3. The regulator ofclaim 1, wherein the first sub-controller comprises a feedforwardcontroller.
 4. The regulator of claim 1, wherein the secondsub-controller comprises an error compensator controller.
 5. Theregulator of claim 4, wherein the wherein the second sub-controllercomprises a form of Proportional-Integral-Derivative, PID, controller.6. The regulator of claim 1, wherein the second sub-controller uses a DClevel as a reference.
 7. The regulator of claim 1, wherein the at leastone characteristic comprises one or more of the: voltage; current; andpower of the electrical output.
 8. The regulator of claim 1, wherein thefirst control signal comprises a first current-control signal and afirst voltage-control signal and wherein the second control signalcomprises a second current-control signal and a second voltage-controlsignal, the combined control signal comprising: a combinedcurrent-control signal to adjust the electrical output based on thefirst and second current-control signals; and a combined voltage-controlsignal to adjust the electrical output based on the first and secondvoltage-control signals.
 9. The regulator of claim 8, wherein the firstsub-controller comprises: a first current-control sub-controller,configured to provide the first current-control signal on the basis ofthe at least one characteristic; and a first voltage-controlsub-controller, configured to provide a first voltage-control signal onthe basis of the at least one characteristic.
 10. The regulator of claim8, wherein the second sub-controller comprises: a second current-controlsub-controller, configured to provide the second current-control signalon the basis of the at least one characteristic; and a secondvoltage-control sub-controller, configured to provide the secondvoltage-control signal on the basis of the at least one characteristic.11. The regulator of claim 1, wherein each of the first and secondsub-controllers each comprise a respective error signal generator, eacherror signal generator being configured to generate a respective errorsignal by comparing the at least one characteristic with a respectivereference value and being further configured to generate the respectivecontrol signal so as to minimise the respective error signal.
 12. Theregulator of claim 1, wherein the combined control signal comprises atleast one Pulse Width Modulation, PWM, signal.
 13. The regulator ofclaim 12, wherein the first and second control signals are PWM signals.14. The regulator of claim 1, wherein the input comprises terminalsconfigured to receive the electrical output from the electricalgenerator, the regulator further comprising: an electronic load,arranged across the input terminals and configured to receive thecombined control signal from the output and to set its resistance basedon the received combined control signal, to adjust the electrical outputthereby.
 15. The regulator of claim 14, wherein the first control signalcomprises a first current-control signal and a first voltage-controlsignal and wherein the second control signal comprises a secondcurrent-control signal and a second voltage-control signal, the combinedcontrol signal comprising: a combined current-control signal to adjustthe electrical output based on the first and second current-controlsignals; and a combined voltage-control signal to adjust the electricaloutput based on the first and second voltage-control signals; andwherein the electronic load is responsive to the combinedvoltage-control signal.
 16. The regulator of claim 1, wherein the inputcomprises terminals configured to receive the electrical output from theelectrical generator, the regulator further comprising: an AC converter,arranged at the input terminals and configured to receive the combinedcontrol signal from the output and to adjust the electrical output inaccordance with the received combined control signal.
 17. The regulatorof claim 16, wherein the first control signal comprises a firstcurrent-control signal and a first voltage-control signal and whereinthe second control signal comprises a second current-control signal anda second voltage-control signal, the combined control signal comprising:a combined current-control signal to adjust the electrical output basedon the first and second current-control signals; and a combinedvoltage-control signal to adjust the electrical output based on thefirst and second voltage-control signals; and wherein the AC converteris only responsive to the combined current-control signal.
 18. Theregulator of claim 1, wherein the input is configured to receive theelectrical output from the electrical generator and wherein the inputcomprises a signal processing module arranged to generate a secondsignal indicative of the at least one characteristic by processing ofthe electrical output, the first and second sub-controllers beingresponsive to the second signal indicative of the at least onecharacteristic.
 19. A method of regulating the electrical output of agenerator, comprising: receiving a signal indicative of at least onecharacteristic of the electrical output; generating a first controlsignal on the basis of the at least one characteristic using a first,relatively fast-response sub-controller; generating a second controlsignal on the basis of the at least one characteristic using a second,relatively slow-response sub-controller; and adjusting the electricaloutput using a combined control signal that is based on the first andsecond control signals.
 20. A computer program, configured when operatedon a processor to carry out the method of claim 19.